Prior art current resonant switching power source devices are widely well known as beneficially indicating less noise and high power conversion efficiency because they can accomplish a zero current switching (ZCS) with a resonance circuit which converts electric current flowing through a just switched switching element into a sine wave form for reduction in switching loss. By way of an example, a prior art current resonant switching power source shown in FIG. 15, comprises a first MOS-FET 2 of high voltage side as a first switching element and a second MOS-FET 3 of low voltage side as a second switching element, first and second MOS-FETs 2 and 3 being connected in series to a DC power source 1; a first capacitor 6 for voltage resonance connected in parallel to second MOS-FET 3; a second capacitor 4 for current resonance and a primary winding 5a of a transformer 5 connected in parallel to first capacitor 6; a rectifying smoother 9 which includes an output diode 7 for commutation and an output capacitor 8 for smoothing connected between a secondary winding 5b of transformer 5 and DC output terminals 10 and 11; and a control circuit 13 for forwarding drive signals VGH and VGL to first and second MOS-FETs 2 and 3 in response to DC output voltage VRL applied from rectifying smoother 9 through DC output terminals 10 and 11 to an electric load 12. Transformer 5 has an equivalent circuit which includes a leakage inductance 5d connected in series to primary winding 5a and an excitation inductance 5e connected in parallel to primary winding 5a. A resonant series circuit 14 comprises second capacitor 4, primary winding 5a, excitation inductance 5e and leakage inductance 5d of transformer 5 connected in parallel to second MOS-FET 3. In the current resonant switching power source device shown in FIG. 15, control circuit 13 produces drive signals VGH and VGL to alternately turn first and second MOS-FETs 2 and 3 on and off so that charging and discharging currents as resonance current flow through leakage inductance 5d and primary winding 5a into and from second capacitor 4 to supply DC output of substantially constant DC voltage VRL to load 12 from secondary winding 5b of transformer 5 through rectifying smoother 9. Control circuit 13 picks up DC output voltage VRL applied from output capacitor 8 of rectifying smoother 9 through DC output terminals 10 and 11 to load 12; determines the on-period of first MOS-FET 2 depending on level of an error voltage, the differential between detected output voltage VRL and reference voltage for regulating a criterion measure of output voltage VRL; sets the on-period of second MOS-FET 3 for a given period over a half cycle in a sine wave of resonance frequency determined by capacitance in second capacitor 4 and leakage inductance 5d of transformer 5; and executes the alternate on-off operation of first and second MOS-FETs 2 and 3.
FIGS. 16 and 17 indicate a waveform chart of voltages and electric currents at selected locations of the current resonant switching power source device shown in FIG. 15. FIGS. 16 and 17 show the variations during respectively heavy and no or light loads wherein time charts (A) and (B) represent voltage variation in drive signals VGH and VGL applied to each gate terminal of first and second MOS-FETs 2 and 3; (C) and (D) represent respectively voltage VQH applied between drain and source terminals of first MOS-FET 2, and electric current IQH flowing through first MOS-FET 2; (E) and (F) represent respectively voltage VQL applied between drain and source terminals of second MOS-FET 3, and electric current IQL flowing through second MOS-FET 3; (G) represents electric current IDO flowing through output diode 7 of rectifying smoother 9; and (H) represents electric current ILP flowing through excitation inductance 5e of transformer 5. As shown in FIG. 17 (H), excitation current for transformer 5, namely, electric current flowing through excitation inductance 5e of transformer 5, oscillates at a substantially same rate across a zero line toward positive and negative sides during no or light load, however, FIG. 16 (H) shows that a central line of the amplitude is biased toward negative side to unilaterally excite transformer 5 during heavy load. Also, FIGS. 18 (A) and (B) indicate variation in magnetic flux φ of magnetic core in transformer 5 relative to excitation current ILP for transformer 5 under respectively no or light and heavy loads. In detail, FIG. 18 (A) reveals magnetization by magnetic flux at a substantially same rate across a zero line toward positive and negative sides under no or light load, however on the contrary, FIG. 18 (B) exemplifies magnetization by magnetic flux biased toward negative side under heavy load. For explanatory convenience, coersive force and residual flux are neglected in FIGS. 18 (A) and (B).
In the device shown in FIG. 15, when first MOS-FET 2 is turned on during period from point t0 to t1, resonance current runs through resonant series circuit 14 of leakage inductance 5d and excitation inductance 5e of transformer 5 and second capacitor 4 to electric charge second capacitor 4 and simultaneously excite transformer 5. Then, when first MOS-FET 2 is turned off at point t1, electric current flowing through resonant series circuit 14 is diverted and sent through a parasitic diode 3a in second MOS-FET 3. At this time, since almost no voltage VQL is impressed between drain and source terminals of second MOS-FET 3, second MOS-FET 3 can be turned on from off for zero voltage switching (ZVS).
When second MOS-FET 3 is turned on at point t2, second capacitor 4 is discharged by resonance current flowing through resonant series circuit 14 so that resonance current passes from second capacitor 4 through primary winding 5a and leakage inductance 5d of transformer 5 and second MOS-FET 3 to second capacitor 4. At this moment, voltage induced on secondary winding 5b of transformer 5 is clamped with DC output voltage VRL, and therefore, voltage developed on primary winding 5a of transformer 5 is clamped with a product value of DC output voltage VRL and turn ratio NP/NS of transformer 5 (NP and NS: numbers of turn for respectively primary and secondary windings 5a and 5b). In this way, resonance current flows through resonant series circuit 14 due to resonant action by leakage inductance 5d of transformer 5 and second capacitor 4, during which electric energy can be transmitted to secondary side. After point t2, negative excitation current ILP through excitation inductance 5e is gradually reduced due to resonant action by leakage and excitation inductances 5d and 5e of transformer 5 and second capacitor 4, and then, after excitation current ILP exceeds zero level, it begins flowing in the adverse direction because excitation current ILP flows from leakage and excitation inductances 5d and 5e into second capacitor 4. Accordingly, as transformer 5 is excited in the adverse direction, voltage imposed on primary winding 5a of transformer 5 is lowered. When the imposed voltage drops below the product of DC output voltage VRL and turn ratio NP/NS of transformer 5, voltage on primary winding 5a is released from clamping not to transport electric energy to secondary side. During this period, only resonance current flows through resonant series circuit 14 by virtue of resonant action by leakage and excitation inductances 5d and 5e of transformer 5. Thereafter, when second MOS-FET 3 is turned off at point t3, electric current flowing through resonant series circuit 14 is recovered to DC power source 1 through a parasitic diode 2a of first MOS-FET 2. During this period, as almost no voltage is applied on drain and source terminals of first MOS-FET 2, zero voltage switching can be accomplished by switching first MOS-FET 2 from off to on during this period. Repetition of the foregoing operations causes pulsatile voltage to appear on secondary winding 5b of transformer 5 and convert through rectifying smoother 9 into a DC output voltage VRL applied on load 12 connected to DC output terminals 10 and 11.
When load 12 becomes heavier than in usual operation, electric energy transmitted from primary to secondary side of transformer 5 during the on-period of second MOS-FET 3 becomes greater with increase in discharged amount from second capacitor 4, diminishing charged voltage on second capacitor 4. At this moment, first MOS-FET 2 is turned on, voltage applied from DC power source 1 on primary winding 5a rises, and increasing excitation current ILP for transformer 5 flows through resonant series circuit 14 of second capacitor 4, and primary winding 5a and excitation and leakage inductances 5e and 5d of transformer 5, returning charged voltage on second capacitor 4 to the original level to establish stabilized DC output voltage VRL on load 12. Thus, if load 12 becomes heavier than in usual operation, excitation current ILP for transformer 5 increases during the on-period of first MOS-FET 2, the device serves to keep charged voltage on second capacitor 4 at a constant level when second MOS-FET 3 is turned on. For that reason, excitation current ILP for transformer 5 during heavy load has a same amplitude as that during no or light load as shown in FIGS. 18 (A) and 18 (B), but there is an increase in unidirectional electric current, that is, electric current flowing through first MOS-FET 2 turned on.
As mentioned above, the current resonant switching power source device shown in FIG. 15 is designed to charge second capacitor 4 with electric current flowing through resonant series circuit 14 to transmit electric energy accumulated in second capacitor 4 to secondary side. Accordingly, unless magnitude of excitation current for transformer 5 is determined in accordance with energy amount necessary at the time of heavy load, no sufficient amount of energy can be supplied to load 12. On the other hand, the on-period of second MOS-FET 3 is a stationary time span over a half cycle of sine wave having the resonant frequency determined by capacitance of leakage inductance 5a of transformer 5 and second capacitor 4. Then, when second MOS-FET 3 is turned on, electric energy is transmitted to secondary side, and at the same time, transformer 5 is magnetized in the adverse direction. Accordingly, magnetized amount in the adverse direction is constant or same regardless of magnitude in load. In other words, amplitude of excitation current for transformer 5 comes to a constant level independently of any condition of load 12, and a large amount of excitation current flows even during no or light load. Accordingly, power loss in transformer 5 incurred by excitation current is not lowered even during no or light load, thereby causing considerable reduction in power conversion efficiency.
To solve the foregoing problem, a proposal has been made in U.S. Pat. No. 6,418,038 to M. Takahama, et al., which discloses a resonant DC-DC converter of bridge type for setting a switching frequency during light load above resonance frequency determined by an inductance of a primary coil and an interwinding capacitance of a secondary coil in a converter transformer. In detail, during heavy load, the converter is operated with a switching frequency near resonance frequency of resonant series circuit which comprises a resonance inductance, inductance of primary coil in transformer and resonance capacitor, and during light load, the converter is operated to stabilize output voltage with switching frequency near resonance frequency of parallel resonant circuit which comprises resonance inductance, inductance of primary coil in transformer, and equivalent capacitance in primary coil equivalent to the interwinding capacitance in secondary coil. As switching frequency during light load is higher than that during heavy load, excitation current during light load can be reduced accordingly to improve power conversion efficiency.
However, the foregoing DC-DC converter of bridge type has a trouble in normal operation because it does not produce sufficient interwinding capacity in secondary coil under the low secondary output voltage, thereby causing switching frequency during light load to abnormally rise. Also, it disadvantageously has a large fluctuation in switching frequency between light and heavy loads, and this prevents stabilized operation of the converter.
Accordingly, an object of the present invention is to provide a resonant switching power source device capable of reducing power loss during light or no load to improve power conversion efficiency during light load.
Another object of the present invention is to provide a resonant switching power source device capable of performing steady operation in any load condition.